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參數(shù)資料
| 型號： | ML4851 |
| 廠商： | Fairchild Semiconductor Corporation |
| 英文描述： | Low Current, Voltage Boost Regulator(小電流升壓穩(wěn)壓器) |
| 中文描述： | 低電流，電壓升壓穩(wěn)壓器（小電流升壓穩(wěn)壓器） |
| 文件頁數(shù)： | 5/9頁 |
| 文件大?。?/td> | 141K |
| 代理商： | ML4851 |
ML4851
REV. 1.0 10/12/2000
5
FUNCTIONAL DESCRIPTION
The ML4851 combines Pulse Frequency Modulation (PFM)
and synchronous rectification to create a boost converter
that is both highly efficient and simple to use. A PFM
regulator charges a single inductor for a fixed period of
time and then completely discharges before another cycle
begins, simplifying the design by eliminating the need for
conventional current limiting circuitry. Synchronous
rectification is accomplished by replacing an external
Schottky diode with an on-chip PMOS device, reducing
switching losses and external component count.
REGULATOR OPERATION
A block diagram of the boost converter is shown in Figure
2. The circuit remains idle when V
OUT
is at or above the
desired output voltage, drawing 50μA from V
IN
, and 8μA
from V
OUT
through the feedback resistors R1 and R2.
When V
OUT
drops below the desired output level, the
output of amplifier A1 goes high, signaling the regulator
to deliver charge to the output. Since the output of
amplifier A2 is normally high, the flip-flop captures the
A1 set signal and creates a pulse at the gate of the NMOS
transistor Q1. The NMOS transistor will charge the
inductor L1 for 5μs, resulting in a peak current given by:
(1)
For reliable operation, L1 should be chosen so that I
L(PEAK)
does not exceed 1A.
When the one-shot times out, the NMOS transistor
releases the V
L
pin, allowing the inductor to fly-back and
momentarily charge the output through the body diode of
PMOS transistor Q2. But, as the voltage across the PMOS
transistor changes polarity, its gate will be driven low by
the current sense amplifier A2, causing Q2 to short out its
body diode. The inductor then discharges into the load
through Q2. The output of A2 also serves to reset the flip-
flop and one-shot in preparation for the next charging
cycle. A2 releases the gate of Q2 when its current falls to
zero. If V
OUT
is still low, the flip-flop will immediately
initiate another pulse. The output capacitor (C1) filters the
inductor current, limiting output voltage ripple. Inductor
current and one-shot waveforms are shown in Figure 3.
RESET
COMPARATOR
An additional comparator is provided to detect low V
IN
,
or any other error condition that is important to the user.
The inverting input of the comparator is internally
connected to V
REF
, while the non-inverting input is
provided externally at the DETECT pin. The output of the
comparator is the
RESET
pin, which swings from V
OUT
to
GND when an error is detected. (Refer to Block Diagram)
DESIGN CONSIDERATIONS
INDUCTOR
Selecting the proper inductor for a specific application
usually involves a trade-off between efficiency and
maximum output current. Choosing too high a value will
keep the regulator from delivering the required output
current under worst case conditions. Choosing too low a
value causes efficiency to suffer. It is necessary to know
the maximum required output current and the input
voltage range to select the proper inductor value. The
maximum inductor value can be estimated using the
following formula:
×
×
×
2
(2)
where
η
is the efficiency, typically between 0.8 and 0.9.
Note that this is the value of inductance that just barely
delivers the required output current under worst case
conditions. A lower value may be required to cover
inductor tolerance, the effect of lower peak inductor
currents caused by resistive losses, and minimum dead
time between pulses.
Another method of determining the appropriate inductor
value is to make an estimate based on the typical
performance curves given in Figures 4 and 5. Figure 4
shows maximum output current as a function of input
voltage for several inductor values. These are typical
performance curves and leave no margin for inductance
and ON-time variations. To accommodate worst case
conditions, it is necessary to derate these curves by at
least 10% in addition to inductor tolerance.
For example, a two cell to 5V application requires 60mA
of output current while using an inductor with 15%
tolerance. The output current should be derated by 25% to
80mA to cover the combined inductor and ON-time
tolerances. Assuming that 2V is the end of life voltage of
a two cell input, Figure 4 shows that with a 2V input, the
ML4851-5 delivers 80mA with an 18μH inductor.
Figure 5 shows efficiency under the conditions used to
create Figure 4. It can be seen that efficiency is mostly
independent of input voltage and is closely related to
inductor value. This illustrates the need to keep the
inductor value as high as possible to attain peak system
efficiency. As the inductor value goes down to 10μH, the
efficiency drops to around 75%. With 33μH, the
efficiency exceeds 90% and there is little room for
improvement. At values greater than 33μH, the operation
of the synchronous rectifier becomes unreliable at low
input voltages because the inductor current is so small
that it is difficult for the control circuitry to detect. The
data used to generate Figures 4 and 5 is provided in Table
1.
I
t
V
L
s
V
L
L PEAK
(
ON
IN
IN
)
=
×
1
=
×
1
5
μ
L
V
t
V
I
MAX
IN MIN
(
ON MIN
(
OUT
OUT MAX
(
=
×
)
)
)
2
η
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