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參數(shù)資料
| 型號(hào)： | ISL6251AHRZ |
| 廠商： | INTERSIL CORP |
| 元件分類： | 電源管理 |
| 英文描述： | Low Cost Multi-Chemistry Battery Charger Controller |
| 中文描述： | 1-CHANNEL POWER SUPPLY SUPPORT CKT, PQCC28 |
| 封裝： | LEAD FREE, 5 X 5 MM, PLASTIC, QFN-28 |
| 文件頁數(shù)： | 15/20頁 |
| 文件大?。?/td> | 593K |
| 代理商： | ISL6251AHRZ |
15
FN9202.1
June 17, 2005
between 0.53 and 0.88 for the minimum battery voltage of
10V (2.5V/Cell) and the maximum battery voltage of 16.8V.
For V
IN,MAX
=19V, VBAT=16.8V, L=10μH, and f
s
=300kHz,
the maximum RMS current is 0.19A. A typical 10F ceramic
capacitor is a good choice to absorb this current and also
has very small size. The tantalum capacitor has a known
failure mechanism when subjected to high surge current.
EMI considerations usually make it desirable to minimize
ripple current in the battery leads. Beads may be added in
series with the battery pack to increase the battery
impedance at 300kHz switching frequency. Switching ripple
current splits between the battery and the output capacitor
depending on the ESR of the output capacitor and battery
impedance. If the ESR of the output capacitor is 10m
and
battery impedance is raised to 2
with a bead, then only
0.5% of the ripple current will flow in the battery.
MOSFET Selection
The Notebook battery charger synchronous buck converter
has the input voltage from the AC adapter output. The
maximum AC adapter output voltage does not exceed 25V.
Therefore, 30V logic MOSFET should be used.
The high side MOSFET must be able to dissipate the
conduction losses plus the switching losses. For the battery
charger application, the input voltage of the synchronous
buck converter is equal to the AC adapter output voltage,
which is relatively constant. The maximum efficiency is
achieved by selecting a high side MOSFET that has the
conduction losses equal to the switching losses. Ensure that
ISL6251 LGATE gate driver can supply sufficient gate
current to prevent it from conduction, which is due to the
injected current into the drain-to-source parasitic capacitor
(Miller capacitor C
gd
), and caused by the voltage rising rate
at phase node at the time instant of the high-side MOSFET
turning on; otherwise, cross-conduction problems may
occur. Reasonably slowing turn-on speed of the high-side
MOSFET by connecting a resistor between the BOOT pin
and gate drive supply source, and the high sink current
capability of the low-side MOSFET gate driver help reduce
the possibility of cross-conduction.
For the high-side MOSFET, the worst-case conduction
losses occur at the minimum input voltage:
The optimum efficiency occurs when the switching losses
equal the conduction losses. However, it is difficult to
calculate the switching losses in the high-side MOSFET
since it must allow for difficult-to-quantify factors that
influence the turn-on and turn-off times. These factors
include the MOSFET internal gate resistance, gate charge,
threshold voltage, stray inductance, pull-up and pull-down
resistance of the gate driver. The following switching loss
calculation provides a rough estimate.
Where Q
gd
: drain-to-gate charge, Q
rr
: total reverse recovery
charge of the body-diode in low side MOSFET, I
LV
: inductor
valley current, I
LP:
Inductor peak current, I
g,sink
and
I
g
,
source
are the peak gate-drive source/sink current of Q1,
respectively.
To achieve low switching losses, it requires low drain-to-gate
charge Q
gd
. Generally, the lower the drain-to-gate charge,
the higher the on-resistance. Therefore, there is a trade-off
between the on-resistance and drain-to-gate charge. Good
MOSFET selection is based on the Figure of Merit (FOM),
which is a product of the total gate charge and on-
resistance. Usually, the smaller the value of FOM, the higher
the efficiency for the same application.
For the low-side MOSFET, the worst-case power dissipation
occurs at minimum battery voltage and maximum input
voltage:
Choose a low-side MOSFET that has the lowest possible
on-resistance with a moderate-sized package like the SO-8
and is reasonably priced. The switching losses are not an
issue for the low side MOSFET because it operates at zero-
voltage-switching.
Choose a Schottky diode in parallel with low-side MOSFET
Q2 with a forward voltage drop low enough to prevent the
low-side MOSFET Q2 body-diode from turning on during the
dead time. This also reduces the power loss in the high-side
MOSFET associated with the reverse recovery of the low-
side MOSFET Q2 body diode.
As a general rule, select a diode with DC current rating equal
to one-third of the load current. One option is to choose a
combined MOSFET with the Schottky diode in a single
package. The integrated packages may work better in
practice because there is less stray inductance due to a
short connection. This Schottky diode is optional and may be
removed if efficiency loss can be tolerated. In addition,
ensure that the required total gate drive current for the
selected MOSFETs should be less than 24mA. So, the total
gate charge for the high-side and low-side MOSFETs is
limited by the following equation:
Where I
GATE
is the total gate drive current and should be
less than 24mA. Substituting I
GATE
=24mA and f
s
=300kHz
into the above equation yields that the total gate charge
DSON
2
BAT
IN
OUT
V
Conduction
,
Q
P
R
I
V
=
s
IN
rr
k
sin
,
g
gd
s
LP
IN
source
,
g
gd
s
LV
IN
Switching
,
Q
P
f
V
Q
I
Q
f
I
V
2
1
I
Q
f
I
V
2
1
+
+
=
DSON
2
BAT
IN
OUT
V
2
Q
R
I
V
1
P
=
s
GATE
f
GATE
Q
I
≤
ISL6251, ISL6251A
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